1. Technical Field
This invention relates generally to a power supply, more particularly, to a self-oscillating switching power supply (SOP) adapted to supply DC current to wide range of loads.
2. Related Art
A switch-mode power supply (SMPS) can operate in or between two current-conduction modes, continuous conduction mode (CCM), and discontinuous conduction mode (DCM). By controlling the power switch with a flyback (feedback signal) to monitor the energy (e.g., current) remaining in an inductor coil, a self-oscillating switching power supply (SOP) can operate at the critical-conduction point between the continuous and discontinuous conduction modes, wherein the power supply begins a new switching cycle at the exact point in time when an output-current inductor coil""s (e.g., a transformer""s secondary coil""s) current (i.e., energy) falls to zero (i.e., approaches zero or is zero). A self-oscillating (flyback-driven) switching power supply (SOP) will include an input-current inductor coil and an output-current inductor coil, but may be implemented with or without a transformer. In a transformer-less (i.e., no transformer) SOP, the input-current inductor coil will also be the output-current inductor coil (e.g., a there will be only a single inductor coil for energy input and output).
FIG. 1A is circuit diagram depicting a typical topology of a transformer (T1)-based self-oscillating (i.e., flyback) switching power supply (SOP) 100 of the related art. The SOP 100 includes a power switch SW1 for interrupting a current I1 through an input-current inductor coil (e.g., primary winding L1 of transformer T1). The power switch SW1 may be implemented as a metal oxide semiconductor field effect transistor (MOSFET) or a insulated gate bipolar transistor (IGBT), or a mechanical switch, etc, or by any suitable presently know or future electrical current-switching device. The power switch SW1 has two states, an xe2x80x9cONxe2x80x9d state characterized by a low impedance, and an xe2x80x9cOFFxe2x80x9d state characterized by a high impedance. The power switch SW1 is generally cyclically turned ON and OFF in a periodic manner, such that the power switch SW1 is ON during a first xe2x80x9cON-timexe2x80x9d period and then OFF during a first xe2x80x9cOFF-timexe2x80x9d period, and then ON again during a second xe2x80x9cON-timexe2x80x9d period (tON) and then OFF during a second xe2x80x9cOFF-timexe2x80x9d period (tOFF), and so forth. The switching frequency FSW of the SOP 100 is calculated as the inverse of the sum of the ON-time plus the subsequent OFF-time (i.e., FSW=1/(xe2x80x9cON-timexe2x80x9d+xe2x80x9cOFF-timexe2x80x9d). The duty cycle (QS) of the SOP 100 is calculated as the ratio of the ON-time to the sum of the ON-time plus the subsequent OFF-time (i.e., QS=xe2x80x9cON-timexe2x80x9d/(xe2x80x9cON-timexe2x80x9d+xe2x80x9cOFF-timexe2x80x9d)).
In general, because there is inductive energy storage in the SOP 100, and a capacitance associated with the GATE terminal of the power switch SW1, a xe2x80x9cminimum ON-timexe2x80x9d (tONMIN) will be characterized by the characteristics of the power switch SW1 and other characteristics of the SOP 100. During normal operation (e.g., critical conduction mode operation) of the SOP 100, the OFF-time will be characterized by (and equal to) the time it takes for the current (i.e., energy) in an output inductor coil (e.g., a transformer secondary coil L2 and/or transformer auxiliary secondary coil L3) to fall to zero (i.e., to approach zero or to be zero). During any discontinuous conduction mode (DCM) operation of the SOP 100, the OFF-time will be longer than the time it takes for the current (i.e., energy) in an output inductor coil (e.g., a transformer secondary coil L2 and/or transformer auxiliary secondary coil L3) to fall to zero. During any continuous conduction mode (CCM) operation of the SOP 100, the OFF-time will be substantially less than the time it would otherwise take for the current (i.e., energy) in an output inductor coil (e.g., a transformer secondary coil L2 and/or transformer auxiliary secondary coil L3) to fall to zero, and the current will not fall all the way to zero.
The power switch SW1 is gated (i.e. controlled ON and OFF) by a switch-control signal asserted on the GATE node of the power switch SW1 by a switch-driver circuit, such as the Frequency Clamped Flyback Driver 110. Frequency clamped flyback switch driver 110 can be implemented with an integrated circuit chip manufactured by Motorola Corp. known as an MC33364 critical-conduction mode controller chip (See, e.g., FIG. 1C).
The power switch SW1 alternately opens and closes, alternately passing and interrupting an input current (I1) which is driven through the transformer""s primary coil L1 by the voltage potential difference (V1) between a power source input voltage VIN and power switch SW1. (In most real circuits, the ON resistance of the power switch SW1 will be negligibly small, such that V1 is approximately equal to voltage VIN when the power switch SW1 is closed). Power source voltage VIN may be a fixed DC voltage or a variable DC voltage (e.g., a DC voltage having a ripple due to lack of filtering of a rectified AC). Persons skilled in the art will recognize that VIN may be provided as a substantially direct current (DC) voltage produced from an alternating current (AC) input voltage (i.e., a line voltage) source via a diode bridge rectifier (not shown) that full-wave rectifies the alternating current and a filter capacitor (not shown) that filters and smooths current pulses received from the bridge rectifier.
The SOP 100 includes an input-current inductor coil (e.g., primary winding L1) connected in series with a power switch SW1 and between a power source (VIN) and a reference potential (ground). As is commonly known, closing and opening of the power switch SW1 causes energy to be stored as a magnetic field in the input-current inductor coil (e.g., in the primary winding L1) which is transferred to an output-current inductor coil (e.g, the magnetically coupled secondary winding L2) and thereupon output substantially as an output current (I2) driven at a secondary voltage V2 and dissipated through a load associated with an impedance, and/or with a resistance (RLOADEQ). A very small, (i.e., negligible) amount of the input energy is output as an auxiliary output current (IAUX) and dissipated through a sensing circuit within or operatively coupled to the switch-driver circuit (e.g, 110). Because the transformer-based SOP 100 operates by transferring energy between the primary and secondary windings L1 and L2, the turns ratio NT of the windings L1 and L2 may be adjusted to either increase or decrease the output voltage (VOUT) associated with the power source VIN, as needed for a particular application. A rectifier diode D1, and a filter capacitor C1 are connected to output-current inductor coil (e.g, secondary winding L2) as shown in FIG. 1A. The rectifier diode D1 rectifies the current pulses (I2) provided by the output-current inductor coil (e.g, secondary winding L2) and the filter capacitor C2 filters and smooths the rectified current pulses to form a substantially direct current (DC) output voltage VOUT.
The transformer T1 includes a primary winding (L1) (connected in series to the power switch SW1), and at least one secondary winding (e.g., L2 and/or L3). A first secondary winding L2 is provided to output at voltage V2 all, or substantially all, of the energy input to the transformer T1 (e.g., energy input as current I1 in the primary winding L1 at voltage V1). The voltages V1 and V2 are generally related by the equation V2=NT*V1. An auxiliary secondary winding L3 is provided to output, at voltage VAUX, a very small portion, (i.e., a negligible amount or none) of the energy input to the transformer T1. The voltage VAUX across the auxiliary secondary winding L3 is related to the voltage across the first secondary winding L2 by the ratio of turns in each of coils L2 and L3 (when the current I2 is decaying in coil L2). Therefore, voltage VAUX is a fixed proportion of V2 (when a current is flowing in coil L2). When the power switch SW1 is OFF (e.g., following an ON-time), and while energy is being dissipated as a decaying (but non-zero) current I2 in the first secondary winding L2, the voltage VAUX will be non-zero. The magnitude of the voltage VAUX will approach zero (or be zero) at the moment that the current I2 falls to zero. At that moment, during normal (critical conduction mode) operation of the SOP 100, power switch SW1 will be closed, and thereafter, the voltage VAUX will be affected by the voltage V1 and the current I1 across the primary winding L1.
By using the voltage VAUX, and/or associated current IAUX as a feedback (i.e., flyback) signal to the switch-driver circuit (e.g., 110), the SOP 100 can operate in critical conduction mode, wherein the next conduction (i.e., ON-time) of the next cycle is initiated by a Zero Current Detector operatively connected to the auxiliary winding L3. The Zero Current Detector is a circuit for detecting (or anticipating) the occurrence of a zero-current condition in the output-current inductor coil (e.g, secondary winding L2). Various other known alternative methods and circuits for detecting the zero-current condition of the output current I2 in the output-current inductor (L2) can be substituted in the SOP 100 to sustain critical conduction mode operation.
As is understood by persons skilled in the art, the conduction (i.e., ON-time) of current (I1), of each cycle, is terminated when the peak inductor current I1 reaches a threshold level (ITH), as performed by circuits known to persons skilled in the art. The threshold level ITH may be dynamically varied (e.g., for power factor correction) by use of a Multiplier output for comparing to a feedback signal commensurate with the current I1. The Zero Current Detector within (as shown) or associated with or connected to the switch driver (e.g., 110) may indirectly sense a zero-current condition of current I2 in the output-current inductor coil (e.g, secondary winding L2) by monitoring an auxiliary voltage VAUX across a magnetically coupled coil (e.g., auxiliary winding L3).
FIG. 1B is a timing diagram depicting currents and voltages in the SOP 100 of FIG. 1A while operating for several cycles in critical conduction mode. FIG. 1B illustrates the method of the critical-conduction mode operation of the SOP 100 of FIG. 1A. FIG. 1B shows the general shape of the currents I1 and I2 flowing through the coils L1 and L2, and of feedback voltage VAUX during a few representative cycles. When the power switch (SW1 of FIG. 1A) closes, a voltage V1 (i.e., V1 is approximately equal to voltage VIN) is asserted across coil L1 and current I1 ramps up (from zero at the end of the previous cycle), until a threshold current magnitude ITH is reached. During this first ON-time (tON1), a magnetic field builds up in the core (e.g., TCORE) of the input current inductor (e.g., the core of coil L1, which is shared with coils L2 and L3). When the power switch SW1 opens, which begins the OFF-time (tOFF1), the magnetic field collapses, and, according to Lenz""s law, the voltage V1 across the input-current inductor (L1) reverses. In this case, the current (I1) has to find some way to continue its flow and begin its decreasexe2x80x94for example, as current I2 by magnetic coupling through the core TCORE of transformer T1.
Time tAUX is the time required for the current I2 in the output-current inductor coil L2 to fall to zero, sensed as the voltage VAUX across the auxiliary secondary coil L3 approaching zero. If the power switch SW1 turns ON again during the ramp-down phase and before the current I2 reaches zero (i.e., tOFF is less than tAUX), then the supply 100 is operating in continuous-conduction mode (CCM). Alternatively, if the energy-storage capability of the input-current inductor coil (e.g., L1) is such that its magnetically coupled current I1/I2 dries out to zero during the switch""s OFF-time (i.e., tOFF greater than tAUX), the supply 100 is operating in discontinuous-conduction mode (DCM). The amount of xe2x80x9cdead timexe2x80x9d (the difference that tOFF exceeds tAUX) for which the magnetically coupled current I1/I2 stays at a null level defines how strongly the supply 100 operates in DCM. If the current through the coil L2 reaches zero and the power switch SW1 turns on immediately (no dead time), the supply 100 operates in critical-conduction mode.
The operation of the supply 100 is comparable to someone (a bucket operator 110) filling a bucket (transformer T1) with water (electrical current) and then flushing the water into a pressurized water tank (capacitor C1) through a check-valve (D1). The water (current) flows down (as current I1) into the bucket (T1) from a source (VIN), and is flushed out (as current I2) under (higher or lower) pressure V2. In this analogy, the bucket operator (110) first presents the bucket (transformer T1) to the source (ON-time) until its inner level (magnetically coupled current I1/I2) reaches a defined limit. Then, the bucket operator (110) removes the bucket (T1) from the spring (OFF-time) and flushes the water (as current I2) into a tank (C1) that supplies a fire hose nozzle (at VOUT). The bucket (T1) can be totally empty (i.e., zero magnetically coupled current I1/I2) before refilling (DCM), or some water (e.g., current I2) can remain in the bucket (T1) before the user presents the bucket (T1) back to the spring (CCM). Suppose that the bucket operator (110) is skillful such that at each cycle he presents the bucket (T1) to the source (VIN) at the precise instant that the water in the bucket (T1) from the previous cycle is completely flushed (thus operating in critical conduction mode).
The end user is a firefighter (such as a brave fireman of the New York City Fire Department, NYFD) who provides the feedback to the bucket operator (110) via his voice, shouting for more or less flow into the pressure tank (C1). If the flames increases, the firefighter applies water faster (higher load) and requires more pressurized water from the tank (C1) and thus asks the bucket operator (110) to provide the bucket (transformer T1), and therefore the tank (C1), with a higher flow. In other words, the bucket operator (110) will fills his container (T1) longer (ON-time increases).
If the flames decreases, the firefighter requires less pressurized water from the water tank (C1) and thus asks the bucket operator (110) to conserve water (conserve energy) by reducing the flow to the bucket (transformer T1), and thereby reducing the flow into the tank (C1). By reducing the filling time (ON-time) during which the flow (current I1) from the source VIN is filling the bucket (T1), the flushing time (OFF-time) required to flush the water (as current I2) into the tank (C1) is reduced. Thus, while the critical conduction mode is maintained, the cycle period (ON-time plus OFF-time) is reduced, thereby increasing the switching frequency FSW. (Note: Increased switching frequency FSW is associated with increased switching losses, that is, the higher the switching frequency FSW, the more energy is wasted, e.g., as heat in the power switch SW1.) The bucket operator (110) of the related art is very strict about limiting the maximum switching frequency, and he (110) clamps the switching frequency FSW to a predetermined maximum value FSWCLAMPED, and the filling/flushing process leaves the critical conduction (flyback-SOP) mode of operation and remains in DCM at that predetermined fixed frequency FSWCLAMPED. There is a generally practical limit on how quickly the bucket operator (110) can cut (turn-OFF) the flow of current (I1) into the bucket (T1) from the source VIN, and thus there is a minimum time (tONMIN is the shortest practical ON-time) that filling the bucket (T1) can be performed.
In the frequency clamped SOP 100 of the related art, the predetermined maximum frequency FSWCLAMPED of the switching frequency (FSW), is predetermined by the minimum ON-time (tONMIN) and by a fixed predetermined minimum OFF-time (tOFFCLAMPED, e.g., where tOFFCLAMPED is fixed by an RC time-constant circuit including resistor RFREQCLAMP and capacitor CFREQCLAMP, as shown in FIG. 1A). In the Frequency Clamped SOP 100 of the related art, the OFF-time is clamped to a predetermined minimum value which remains fixed by a time-constant circuit (e.g., RFREQCLAMP and CFREQCLAMP of FIG. 1A) even while the load approaches zero or becomes zero (i.e., the load is deemed zero when zero current is output from the power supply 100). The switching frequency FSW is clamped to a predetermined fixed frequency FSWCLAMPED in this manner in order to prevent the switching frequency FSW from shifting to a high value, which otherwise can happen in the absence of a load (i.e., zero load). When operating below this predetermined fixed frequency FSWCLAMPED, the SOP 100 operates in critical conduction mode, with a varying frequency but generally constant duty-cycle. The power (P) consumed by the SOP 100 operating in critical conduction mode is governed by the equation:
P=0.5*FSW*(tON{circumflex over ( )}2)*VIN2/L, 
where tON is the ON-time; and L is the primary inductance of the transformer. When operating at the predetermined fixed frequency FSWCLAMPED, the SOP 100 operates in a discontinuous conduction mode (DCM) with a generally constant duty-cycle. The power (P) consumed by the SOP 100 operating in discontinuous conduction mode (DCM) at the predetermined fixed frequency FSWCLAMPED is governed by the following equation:
P=0.5*FSWCLAMPED*(tONMIN{circumflex over ( )}2)*VIN2/L, 
where FSWCLAMPED=1/(tON+tOFF), and tON=tONMIN, and tOFF is the fixed OFF-time tOFFCLAMPED.
FIG. 1C is a block diagram depicting the internal functions of the Frequency Clamped Flyback Driver 110 of FIG. 1A. The functions of the Frequency Clamped Flyback Driver 110 of FIG. 1C may be performed by a Motorola Corp. switch driver chip known as an MC33368 controller. A switch driver 110 for controlling the ON/OFF state of the power switch SW1 (of FIG. 1A) may include a flip-flop (e.g., set-dominant latch) 118 and a combinatorial logic gate (e.g., NOR-gate 112) connected as shown in FIG. 1A. The combinatorial logic gate (112) combines control signals (e.g., latched zero-current detection signals from the latch 118, and minimum OFF-time signals from the frequency clamp 116) to effectively control the power switch (SW1 of FIG. 1A). Effective control of the power switch SW1 (for critical conduction mode operation) includes closing the switch SW1 immediately upon the occurrence of a zero-current condition of the output-current inductor coil (L2 of the SOP of FIG. 1A), which can be effected by outputting a control signal from the Zero Current Detector (130) to the combinatorial logic gate (112) and resetting the flip-flop 118. The flip-flop (118) latches the (switch-ON) control signal from the Zero-Current Detector (130) so that the switch will stay closed (ON) until the current (I1) through the input-current inductor coil (L1) reaches a threshold current magnitude ITH or until an Output Overvoltage or other undesired condition is detected. Accordingly, the power switch (SW1 of FIG. 1A) will thereafter remain closed (ON) until: the current (I1) through the input-current inductor coil (L1) reaches a threshold current magnitude ITH, the value of ITH being determined by the external Mult control signal into a Multiplier (134); or until an Output Overvoltage is detected by circuit 132 based on feedback signal FB (from the output VOUT of the SOP 100 of FIG. 1A). After the power switch (SW1) opens (i.e., turns OFF), it will be closed again immediately upon the occurrence of a zero-current condition of the output-current inductor coil (L2 of the SOP of FIG. 1A), if and only if the OFF-time associated with switching frequency FSW that would occur during critical conduction mode operation is equal to or greater than the OFF-time associated with a predetermined fixed frequency FSWCLAMPED. Regardless of the magnitude of the load (e.g., no matter how small the magnitude of the current out of the power supply), if the switching frequency FSW that would otherwise occur during critical conduction mode operation is greater than the predetermined fixed frequency FSWCLAMPED, then the switching frequency FSW will be clamped to the predetermined fixed frequency FSWCLAMPED. The Frequency Clamp 116, when controlled (via the Frequency Clamp Pin) by a time-constant circuit (e.g., RFREQCLAMP and CFRQCLAMP of FIG. 1A), will clamp the switching frequency FSW to the predetermined fixed frequency FSWCLAMPED by delaying the termination of the OFF-time (i.e., termination of the OFF-time is commencement of the ON-time) that otherwise would be effected by a zero-current-indicating signal from the Zero-Current Detector (130). An optional amplifying buffer 114 provides amplification of the current and/or voltage necessary to rapidly gate (i.e., turn ON/OFF) the power switch (SW1 of FIG. 1A).
FIG. 1D is a graph depicting the general relationship of switching frequency (FSW) to a (slowly changing) load (e.g., a current ISUP out of the power supply indicated generally by VOUT/RLOADEQ) during medium-load and small-load conditions of the SOP 100 of FIG. 1A. The graph (FIG. 1D) is not drawn to scale, and the scaling factor xe2x80x9cxxe2x80x9d in each abscissa point along the logarithmic RLOADEQ axis depends upon the designed power-rating of a particular power supply made in accordance with the topology of SOP (100) of the related art. As illustrated in FIG. 1D, the SOP 100 of FIG. 1A will operate in critical conduction mode, with switching frequency (FSW) increasing as the load decreases, until the switching frequency (FSW) in critical conduction mode would exceed the predetermined fixed frequency FSWCLAMPED (due to small-load conditions), and FSW is thereafter clamped to the predetermined fixed frequency FSWCLAMPED. The operation of the SOP (100 of FIG. 1A) under small load conditions, including zero-load conditions, incurs avoidable switching losses and consumes unnecessary electrical power.
The present invention overcomes the disadvantages of the self-oscillating switching power supply (SOP) 100 of the related art.
In a first aspect, the present invention provides a switching power supply having an output voltage (VOUT) for supplying a supply current to a dynamically variable load, the switching power supply comprising:
an input-current inductor coil connected in series with an input voltage source (VIN) and a power switch;
an output-current inductor coil for outputting an output current at at least the output voltage (VOUT);
a zero-current detector having a detector-output being activated when the output current falls to zero;
a switch-driver circuit for closing and opening the power switch, the switch-driver circuit including:
a flip-flop adapted to latch the activated detector-output and having a first flip-flop input connected to the detector-output, and having a flip-flop output, the flip-flop output being activated while the activated detector-output is latched;
a combinatorial logic gate, having a first logic gate input connected to the flip-flop output and a second logic gate input and a logic gate output, the logic gate output being activated if the flip-flop output is activated and if the second logic gate input is not inhibited, the power switch being closed while the logic gate output is activated;
a pulse generator having a pulse generator-output for outputting an OFF-pulse having a dynamically variable load-modulated pulse width that corresponds to the dynamically variable load, the pulse generator-output being connected to the second logic gate input and inhibiting the second logic gate input during the pulse-width of the load-modulated OFF-pulse.
In a second aspect, the present invention provides a method for operating a switching power supply adapted to supply a load current through a dynamically variable load, the method comprising:
operating the supply in discontinuous current mode (DCM) while the supply is supplying load current having a first load current magnitude, and increasing the extent (EDCM) that the supply operates in DCM as the load current decreases from the first load current magnitude; and
operating the supply in critical current mode while the supply is supplying load current having a second load current magnitude that is larger than the first load current magnitude.
In a third aspect, the present invention provides a switching power supply adapted to operate in a critical conduction mode and in a discontinuous conduction mode, comprising:
a power switch connected in series to an input-current inductor coil and operatively coupled to an output-current inductor coil, the power switch adapted to interrupt an input current through the input-current inductor coil, the power switch being either in an OFF or an ON state and conducting the input current when in its ON state;
a switching controller adapted to control the state of the power switch, the switching controller being operatively coupled to:
a first feedback signal for indicating a zero-current condition in the output-current inductor coil, wherein the switching controller turns the power switch ON in response to the first feedback signal while the supply operates in the critical conduction mode;
a second feedback signal indicating a threshold current magnitude in the input-current inductor coil, wherein the switching controller turns the power switch OFF in response to the second feedback signal; and
a third feedback signal, wherein the switching controller holds the power switch OFF notwithstanding the first feedback signal during a dynamically variable OFF-time that is varied in response to the third feedback signal while the supply operates in the discontinuous conduction mode.
The foregoing and other features and advantages of the present invention will be apparent from the following description of embodiments of the present invention.